Cascode circuit

ABSTRACT

A cascode circuit for a high-gain or high-output millimeter-wave device that operates with stability. The cascode circuit including two cascode-connected transistors includes: a first high electron mobility transistor (HEMT) including a source that is grounded; a second HEMT including a source connected to a drain of the first HEMT; a reflection gain restricting resistance connected to the gate of the second HEMT, for restricting reflection gain; and an open stub connected to a side of the reflection gain restricting resistance which is opposite the side connected to the second HEMT, for short-circuiting high-frequency signals at a predetermined frequency and nearby frequencies.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a cascode circuit used at amillimeter-wave band.

2. Description of the Related Art

In recent years, the range of application using radio waves at themillimeter-wave band such as a 60 GHz band wireless personal areanetwork (WPAN) and a 76 GHz band millimeter-wave radar is widened. Inassociation with this, a millimeter-wave device is required to have highgain and high output.

A generally known method to improve power gain is to cascode-connecttransistors. Cascode connection is to connect to a drain of asource-grounded transistor a gate-grounded transistor. A circuit formedin this way is referred to as a cascode circuit.

A typical conventional cascode circuit is described in the followingwith reference to the attached drawings.

FIG. 24 is a circuit diagram illustrating a conventional cascodecircuit.

In FIG. 24, a drain of a first transistor 51 a source of which isgrounded is connected to a source of a second transistor 52 a gate ofwhich is grounded. In order to ground a high-frequency signal, a gate ofthe second transistor 52 is grounded through a MIM capacitor 53. A gateof the first transistor 51 is connected to an input terminal while adrain of the second transistor 52 is connected to an output terminal.

It is to be noted that, although the cascode circuit illustrated in FIG.24 is described in the context that high electron mobility transistors(HEMTs) are used as the transistors, the same can be said in a case inwhich hetero-junction bipolar transistors (HBTs) or the like are used.In this case, a base-grounded transistor is connected to a collector ofan emitter-grounded transistor. In the following, a drain, a gate, and asource of an HEMT shall be able to be replaced by a collector, a base,and an emitter of an HBT, respectively.

As described above, a high-frequency signal at the gate of the secondtransistor 52 is grounded through the MIM capacitor 53. However, at themillimeter-wave band, the inductance of wiring connected to the MIMcapacitor 53 and a parasitic inductance of a via hole can not beneglected. Therefore, a high-frequency signal at a desired frequency isshort-circuited through a parasitic component. Therefore, there is aproblem that, at the millimeter-wave band, even if the transistors arecascode-connected, the gain can not be satisfactorily improved.

Here, as an example, frequency characteristics of a maximum availablegain (MAG) in a single emitter-grounded HBT and in a cascode-connectedHBT are illustrated in FIG. 25.

With reference to FIG. 25, for example, at a microwave band of 10 GHz orthe like, by cascode-connecting HBTs, the power gain is larger than thatof the single HBT by about 10 dB. However, at the millimeter-wave band,as the frequency increases, the difference between the power gain of thecascode-connected HBT and the power gain of the single HBT becomessmaller. In particular, at a high-frequency band such as a 60 GHz band,a 76 GHz band, or the like, even a cascode circuit can not obtain asatisfactory gain.

It is to be noted that another method of improving the power gain is tocontinuously connect single transistors in series to increase the gain.However, this method has a problem that, as the number of thetransistors and the number of peripheral circuits increase, the chiparea increases and the cost increases as well.

In order to solve the above-mentioned problems, for example, thefollowing can be referred to.

In a cascode circuit described in Japanese Patent Application Laid-openNo. 2002-359530, as illustrated in FIG. 26, a first transistor 51 and asecond transistor 52 are cascode-connected, and an open stub 54 having alength of about ¼ the wavelength of a signal at an operating frequencyis connected to a gate of the second transistor 52.

Here, because, at the operating frequency, the gate of the secondtransistor 52 is grounded at a high frequency by the open stub 54,compared with a case in which a MIM capacitor and a via hole are formedin the vicinity of the gate and grounding is carried out, the parasiticcomponent has less influence, and thus, satisfactory grounding isenabled.

Therefore, at the operating frequency, compared with a case in which thegrounding is carried out using a MIM capacitor and a via hole, the powergain can be improved.

However, the conventional art has the following problems.

In the conventional cascode circuit disclosed in Japanese PatentApplication Laid-open No. 2002-359530, a reflection gain is caused on anoutput side.

Here, as an example, FIG. 27 illustrates frequency characteristics of areflection gain/loss on an output side in a single HEMT, a cascodecircuit in which a gate of a second transistor is grounded through a MIMcapacitor (see FIG. 24), and a cascode circuit disclosed in JapanesePatent Application Laid-open No. 2002-359530 in which a gate of a secondtransistor is grounded through an open stub (see FIG. 26).

In FIG. 27, the cascode circuit grounded through a MIM capacitor has areflection gain at a frequency band of about 20-90 GHz. The cascodecircuit grounded through an open stub has a reflection gain at afrequency band of about 70 GHz or more.

When the cascode circuit has a reflection gain, unnecessary oscillationis caused. If the cascode circuit is applied to, for example, anamplifier, there is a problem that stable normal operation may not beexpected.

Further, if the cascode circuit is applied to, for example, anoscillator, there is a problem that satisfactory output may not beexpected.

SUMMARY OF THE INVENTION

The present invention has been made to solve the above-mentionedproblems, and an object of the present invention is to provide a cascodecircuit which can realize a high-gain or high-output millimeter-wavedevice that operates with stability at the millimeter-wave band.

According to the present invention, a cascode circuit including twocascode-connected transistors includes: a first transistor including oneof a source and an emitter being grounded; a second transistor includingone of a source and an emitter being connected to one of a drain and acollector of the first transistor; a signal improving circuit connectedto one of a gate and a base of the second transistor, for improving andoutputting an input signal; and a filter circuit connected to a side ofthe signal improving circuit which is opposite to the second transistor,for short-circuiting high-frequency signals in a predetermined frequencyincluding the vicinity of the frequency.

In the cascode circuit according to the present invention, the twotransistors are cascode-connected, the signal improving circuit isconnected to one of the gate and the base of the second transistor, andthe filter circuit is connected to a side of the signal improvingcircuit which is opposite to the second transistor. Here, the signalimproving circuit improves an input signal and outputs the signal.

Therefore, by using the cascode circuit, a high-gain or high-outputmillimeter-wave device can be realized which operates with stability atthe millimeter-wave band.

BRIEF DESCRIPTION OF THE DRAWINGS

In the accompanying drawings:

FIG. 1 is a circuit diagram illustrating a cascode circuit according toEmbodiment 1 of the present invention;

FIG. 2 is an explanatory graph of frequency characteristics of a MAG inthe cascode circuit illustrated in FIG. 1 in comparison with a case of asingle HEMT;

FIG. 3 is an explanatory graph of frequency characteristics of areflection gain/loss on an output side in the cascode circuitillustrated in FIG. 1;

FIG. 4 is a circuit diagram illustrating a structure in which a selectorswitch is used to select an open stub;

FIG. 5 is a circuit diagram illustrating a variable length stub;

FIG. 6 is a circuit diagram illustrating an amplifier according toEmbodiment 2 of the present invention;

FIG. 7 is an explanatory graph of frequency characteristics of a gain inthe amplifier illustrated in FIG. 6;

FIG. 8 is an explanatory graph of frequency characteristics of areflection gain/loss on an input side and on an output side in theamplifier illustrated in FIG. 6;

FIG. 9 is a circuit diagram illustrating an amplifier according toEmbodiment 3 of the present invention;

FIG. 10 is an explanatory graph of frequency characteristics of a MAG inthe amplifier illustrated in FIG. 9;

FIG. 11 is an explanatory graph of frequency characteristics of a gainin the amplifier illustrated in FIG. 9;

FIG. 12 is an explanatory graph of frequency characteristics of areflection gain/loss on an input side and on an output side in theamplifier illustrated in FIG. 9;

FIG. 13 is a circuit diagram illustrating an oscillator according toEmbodiment 4 of the present invention;

FIG. 14 is an explanatory graph of frequency characteristics ofRe(Z_(tr))+Re(Z_(res)) in the oscillator illustrated in FIG. 13;

FIG. 15 is an explanatory graph of frequency characteristics ofIm(Z_(tr))+Im(Z_(res)) in the oscillator illustrated in FIG. 13;

FIG. 16 is an explanatory graph showing a relationship between aresistance ratio of divider resistances and output power in theoscillator illustrated in FIG. 13 with regard to a fundamental wave anda second harmonic of an oscillation signal;

FIG. 17 is an explanatory graph showing a relationship between anelectrical length of a phase adjusting line at an oscillation frequencyand maximum output power of the second harmonic of an oscillation signalin the oscillator illustrated in FIG. 13;

FIG. 18 is a contour map illustrating an output power of the secondharmonic in the oscillator illustrated in FIG. 13 when the electricallength is 18°;

FIG. 19 is an explanatory graph of distribution of harmonics with regardto output power in the oscillator illustrated in FIG. 13;

FIG. 20 is a circuit diagram illustrating an oscillator according toEmbodiment 5 of the present invention;

FIG. 21 is an equivalent circuit diagram of a typical diode;

FIG. 22 is a circuit diagram illustrating another oscillator accordingto Embodiment 5 of the present invention;

FIG. 23 is a circuit diagram illustrating a capacitor for controlling anoscillation frequency of an oscillator;

FIG. 24 is a circuit diagram illustrating a conventional cascodecircuit;

FIG. 25 is an explanatory graph of frequency characteristics of a MAG ina single emitter-grounded HBT and in a cascode-connected HBT;

FIG. 26 is a circuit diagram illustrating another conventional cascodecircuit; and

FIG. 27 is an explanatory graph of frequency characteristics of areflection gain/loss on an output side in a single HEMT, a cascodecircuit in which a gate of a second transistor is grounded through a MIMcapacitor, and a cascode circuit in which a gate of a second transistoris grounded through an open stub.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

With reference to the attached drawings, embodiments of the presentinvention are now described in the following. Throughout the drawings,like reference numerals are used to designate like or identical parts.

Embodiment 1

FIG. 1 is a circuit diagram illustrating a cascode circuit according toEmbodiment 1 of the present invention. It is to be noted that thecascode circuit is formed such that its MAG is optimized at 76 GHz(predetermined frequency).

In FIG. 1, a drain of a source-grounded HEMT 1 (first transistor) isconnected to a source of a gate-grounded HEMT 2 (second transistor). Inother words, the HEMT 1 and the HEMT 2 are cascode-connected. A gate ofthe HEMT 1 is connected to an input terminal and a drain of the HEMT 2is connected to an output terminal.

A gate of the HEMT 2 is connected to a reflection gain restrictingresistance 3 (signal improving circuit) for restricting a reflectiongain. An open stub 4 (filter circuit) for short-circuitinghigh-frequency signals in the predetermined frequency including thevicinity of the frequency is connected to a side of the reflection gainrestricting resistance 3, which is opposite to the HEMT 2. Here, alength of the open stub 4 is set to be shorter than ¼ a wavelength (λ/4)of a high-frequency signal at the predetermined frequency to be used.

Divider resistances 5 and 6 for setting a gate voltage of the HEMT 2 areconnected between a source of the HEMT 1 and the gate of the HEMT 2 andbetween the gate and the drain of the HEMT 2, respectively.

Next, description is made with regard to frequency characteristics of aMAG and a reflection gain/loss in the cascode circuit of theabove-mentioned structure when a resistance value of the reflection gainrestricting resistance 3 is changed as a parameter.

FIG. 2 is an explanatory graph of frequency characteristics of a MAG inthe cascode circuit illustrated in FIG. 1 in comparison with a case of asingle HEMT. FIG. 3 is an explanatory graph of frequency characteristicsof a reflection gain/loss on an output side in the cascode circuitillustrated in FIG. 1.

In FIGS. 2 and 3, when the resistance value of the reflection gainrestricting resistance 3 is 0Ω, although the MAG is the highest, thereflection gain is caused, and thus, the circuit is unstable. When theresistance value of the reflection gain restricting resistance 3 is 40Ω,although the reflection gain is not caused, the MAG is lower than thatin the case of the single HEMT at about 76 GHz, and thus, the advantageof the cascode circuit is not exploited. Here, when the resistance valueof the reflection gain restricting resistance 3 is 20Ω, the reflectiongain is not caused, and at the same time, the MAG is higher than that inthe case of the single HEMT at about 76 GHz.

Therefore, by setting the resistance value of the reflection gainrestricting resistance 3 to be about 20Ω, the reflection gain can berestricted, and at the same time, a satisfactory MAG can be obtained.

Further, by setting the length of the open stub 4 to be shorter thanλ/4, the MAG can be prevented from being lowered due to the reflectiongain restricting resistance 3, that is, a point A indicated in FIG. 2can be shifted to the higher side with respect to the predeterminedfrequency.

It is to be noted that the length of the open stub 4 may be ¼ thewavelength of a high-frequency signal at the predetermined frequency tobe used.

Further, as apparent from FIGS. 2 and 3, at a frequency band of about 65GHz or less, the cascode circuit has a higher MAG than that of thesingle HEMT, and at the same time, the reflection gain is restricted.

Therefore, for example, in an amplifier used at 30 GHz band, the lengthof the open stub does not necessarily have to be λ/4 at the 30 GHz band.More specifically, insofar as the MAG is higher than that in the case ofthe single HEMT and the reflection gain is restricted, the length of theopen stub may be set to be shorter than λ/4 of a high-frequency signalat the predetermined frequency.

In the cascode circuit according to Embodiment 1 of the presentinvention, the first transistor and the second transistor arecascode-connected, and the gate of the second transistor is connected tothe resistance for restricting the reflection gain. The open stub forshort-circuiting high-frequency signals in the predetermined frequencyincluding the vicinity of the frequency is connected to the side of theresistance, which is opposite to the second transistor.

Therefore, a cascode circuit which has a simple structure but stillrestricts the reflection gain and improves the MAG at a millimeter-waveband can be obtained. Further, even when MAGs of the individualtransistors are not satisfactorily high, the cascode connection makes itpossible to improve the MAG.

Further, by using the cascode circuit, a high-gain millimeter-wavedevice can be realized which operates with stability at themillimeter-wave band.

Further, because the structure of the circuit is simple, a chip areadoes not increase and a cost increase can be prevented.

Further, because the frequency band of the MAG is determined not by thecapacitance of a capacitor which varies greatly within a wafer surfacebut by the length of the stub, variations in the characteristics of thecascode circuit are suppressed and the yield can be improved.

It is to be noted that, although, in Embodiment 1, single-gate HEMTs(HEMTs 1 and 2) are cascode-connected to form the cascode circuit, thepresent invention is not limited thereto. For example, in an HEMTprocess, a dual-gate HEMT may be used which is equivalent to asource-grounded HEMT and a gate-grounded HEMT which arecascode-connected to each other.

In this case, the chip area can be decreased.

Further, although, in Embodiment 1, a pair of HEMTs, that is, the HEMT 1and the HEMT 2, are cascode-connected, the present invention is notlimited thereto, and the HEMT 2 may be multistage HEMTs to be connectedto the HEMT 1. Here, the multistage HEMTs may be formed by connecting adrain of an HEMT to sources of other HEMTs one after another in series,or, alternatively, by connecting a drain of an HEMT to sources of aplurality of HEMTs in parallel.

In those cases, the gain and the output can be further improved.

Still further, although, in Embodiment 1, only one open stub 4 is used,the present invention is not limited thereto. As in a cascode circuitillustrated in FIG. 4, a plurality of open stubs may be provided forrespective frequency bands to be used, and an appropriate stub may beselected using a selector switch 41. Further, instead of the open stub4, a variable length stub the length of which can be adjusted by a microelectro mechanical system (MEMS) switch 42 as illustrated in FIG. 5 maybe used to change the frequency band to be used.

In those cases, because the MAG can be made high only with regard to adesired band corresponding to the length of the stub, a gain at anunnecessary band can be easily lowered.

Further, although, conventionally, in a multiband system which uses aplurality of frequency bands, a transistor is necessary for each of thebands, by changing the frequency band using the selector switch 41 orthe variable length stub, even in a multiband system, operation at therespective bands is enabled with a set of cascode circuits. Therefore,the number of necessary transistors is decreased, and the cost can bedecreased.

Further, although, in Embodiment 1, the open stub 4 is used as thefilter circuit for short-circuiting a signal at the predeterminedfrequency, the present invention is not limited thereto. For example, aseries resonant circuit may be used in which an inductor and a capacitorare connected in series so as to produce resonance at the predeterminedfrequency, or a short stub may be used the length of which is ½ thewavelength of a high-frequency signal at the predetermined frequency.

Those cases also have similar effects to those in Embodiment 1.

Embodiment 2

FIG. 6 is a circuit diagram illustrating an amplifier according toEmbodiment 2 of the present invention. It is to be noted that theamplifier is formed such that its gain is the highest at 76 GHz(predetermined frequency).

In FIG. 6, an HEMT 1 and an HEMT 2 are cascode-connected. A gate of theHEMT 1 is connected to an input terminal and a drain of the HEMT 2 isconnected to an output terminal.

A gate of the HEMT 2 is connected to a reflection gain restrictingresistance 3 for restricting a reflection gain. Divider resistances 5and 6 for setting a gate voltage of the HEMT 2 are connected between asource of the HEMT 1 and the gate of the HEMT 2 and between the gate andthe drain of the HEMT 2, respectively.

Instead of the linear open stub 4 illustrated in FIG. 1, a fan-shapedradial stub 7 is connected to a side of the reflection gain restrictingresistance 3, which is opposite to the HEMT 2. Similarly to the openstub 4, a length of the radial stub 7 is set such that a high-frequencysignal at the predetermined frequency is substantially short-circuitedat a node. Further, by using the radial stub 7, a high-frequency signalcan be short-circuited at the node at a wide band.

A gate bias circuit 8 and a matching circuit including an equalizer 9and a coupled line 10 are connected between the gate of the HEMT 1 andthe input terminal. A drain bias circuit 11 and a matching circuitincluding an equalizer 12 and a coupled line 13 are connected betweenthe drain of the HEMT 2 and the output terminal.

The gate bias circuit 8 includes a resistance, a transmission line, acapacitor, and a gate bias terminal, supplies a gate bias, and givesstability to operation of the circuit.

The drain bias circuit 11 includes a short stub having a length of λ/4of a high-frequency signal at the predetermined frequency, a capacitor,and a drain bias terminal, and supplies a drain bias and filters outsignals other than a high-frequency signal at the predeterminedfrequency.

The coupled lines 10 and 13 filter out a direct current signal and asignal at a low-frequency band. The equalizers 9 and 12 lower a gain ata band which can not be filtered out by the coupled lines 10 and 13 andgive stability to operation of the circuit.

In the amplifier structured as described above, a signal input to theinput terminal is amplified and output from the output terminal.

Next, frequency characteristics of a gain and a reflection gain/loss inthe amplifier are described.

FIG. 7 is an explanatory graph of the frequency characteristics of thegain in the amplifier illustrated in FIG. 6. FIG. 8 is an explanatorygraph of the frequency characteristics of the reflection gain/loss on aninput side and on an output side in the amplifier illustrated in FIG. 6.

From FIGS. 7 and 8, it can be seen that, at the predetermined frequencyof 76 GHz, the reflection gain can be restricted and a gain as high asabout 10 dB can be obtained.

In the amplifier according to Embodiment 2 of the present invention, thefirst transistor and the second transistor are cascode-connected, andthe gate of the second transistor is connected to the resistance forrestricting the reflection gain. The radial stub for short-circuitinghigh-frequency signals in the predetermined frequency including thevicinity of the frequency is connected to the side of the resistance,which is opposite to the second transistor.

Therefore, a high-gain amplifier which restricts the reflection gain ata millimeter-wave band can be obtained.

Embodiment 3

FIG. 9 is a circuit diagram illustrating an amplifier according toEmbodiment 3 of the present invention. It is to be noted that theamplifier is formed as a wideband amplifier which can be used over awide band.

In FIG. 9, an HEMT 1 and an HEMT 2 are cascode-connected. A gate of theHEMT 1 is connected to an input terminal and a drain of the HEMT 2 isconnected to an output terminal.

A gate of the HEMT 2 is connected to a reflection gain restrictingresistance 3 for restricting a reflection gain. An open stub 4 forshort-circuiting high-frequency signals in a predetermined frequencyincluding the vicinity of the frequency is connected to a side of thereflection gain restricting resistance 3, which is opposite to the HEMT2. Here, a length of the open stub 4 is set to be shorter than ¼ thewavelength (λ/4) of a high-frequency signal at the predeterminedfrequency to be used (for example, 76 GHz).

Instead of the divider resistance 5 illustrated in FIG. 1, a first diode14 is connected between a source of the HEMT 1 and the gate of the HEMT2. An anode of the first diode 14 is grounded and a cathode of the firstdiode 14 is connected to the gate of the HEMT 2. Further, instead of thedivider resistance 6 illustrated in FIG. 1, a second diode 15 isconnected between the gate and the drain of the HEMT 2. An anode of thesecond diode 15 is connected to the gate of the HEMT 2 and a cathode ofthe second diode 15 is connected to the drain of the HEMT 2.

Because a reverse bias voltage is applied to each of the first diode 14and the second diode 15, the first diode 14 and the second diode 15 canbe regarded as resistances having a high resistance value. Here,ordinarily, the divider resistances are required to have a resistancevalue on the order of several kiloohms. When an epitaxial resistanceobtained by an ordinary gallium arsenide (GaAs) wafer is used, a lengthof the resistance is on the order of several hundred micrometers.

On the other hand, when a Schottky gate of an HEMT is used to form adiode, the diode can be obtained only by short-circuiting a drain and asource of the HEMT.

Therefore, by using a Schottky diode as a divider resistance, a chiparea can be decreased.

It is to be noted that, in the case of, for example, an HBT process, aPN diode between a base and a collector may be used as a dividerresistance.

Further, stabilizing circuits 16 and 17 for making the amplifier usableover a wide band are connected between the gate of the HEMT 1 and thegate of the HEMT 2 and between the gate of the HEMT 2 and the drain ofthe HEMT 2, respectively. Each of the stabilizing circuits 16 and 17 isformed by connecting a resistance and a capacitor in series.

A gate bias circuit 8 and a matching circuit including a capacitor 18are connected between the gate of the HEMT 1 and the input terminal. Adrain bias circuit 11 and a matching circuit including a capacitor 19are connected between the drain of the HEMT 2 and the output terminal.

The gate bias circuit 8 and the drain bias circuit 11 have functionssimilar to those of the gate bias circuit 8 and the drain bias circuit11, respectively, illustrated in FIG. 6. The capacitors 18 and 19filters out a signal at a low-frequency band.

In the amplifier structured as described above, a signal input to theinput terminal is amplified and output from the output terminal.

Next, frequency characteristics of a MAG, a gain, and a reflectiongain/loss in the amplifier are described.

FIG. 10 is an explanatory graph of frequency characteristics of a MAG inthe amplifier illustrated in FIG. 9. FIG. 11 is an explanatory graph offrequency characteristics of a gain in the amplifier illustrated in FIG.9. FIG. 12 is an explanatory graph of frequency characteristics of areflection gain/loss on an input side and on an output side in theamplifier illustrated in FIG. 9.

With reference to FIGS. 10 to 12, by virtue of the open stub 4, the MAGis increased at about 90 GHz, while, by virtue of the stabilizingcircuits 16 and 17, the MAG is flat from about 30 GHz to about 80 GHz.

Here, the MAG has a gain also at frequency bands of 30 GHz or less and90 GHz or more. However, because a signal at the low-frequency band of30 GHz or less is filtered out by the capacitors 18 and 19 and a signalat the high-frequency band of 90 GHz or more is filtered out by thedrain bias circuit 11 having a filtering function, the gain of theamplifier has the characteristics as illustrated in FIG. 11.

Further, from FIGS. 10 to 12, it can be seen that a third band as largeas 30-90 GHz can be secured and that, at this band, the reflection gainis restricted and a gain as high as 5 dB or more is obtained even at themillimeter-wave band.

Although a typical wideband amplifier has a complicated peripheralcircuit such as a pre-matching circuit, the present invention canrealize a wideband amplifier with a relatively simple circuit structureand a relatively simple design.

In the amplifier according to Embodiment 3 of the present invention, thefirst transistor and the second transistor are cascode-connected, andthe gate of the second transistor is connected to the resistance forrestricting the reflection gain. The open stub for short-circuitinghigh-frequency signals in the predetermined frequency including thevicinity of the frequency is connected to the side of the resistancewhich is opposite to the second transistor.

Further, the stabilizing circuits are connected between the gate of thefirst transistor and the gate of the second transistor and between thegate of the second transistor and the drain of the second transistor,respectively. Further, a gate bias circuit and a matching circuit areconnected between the gate of the first transistor and the inputterminal, and another drain bias circuit and another matching circuitare connected between the drain of the second transistor and the outputterminal.

Therefore, a high-gain amplifier which restricts the reflection gainover a wide band at the millimeter-wave band can be obtained.

Embodiment 4

FIG. 13 is a circuit diagram illustrating an oscillator according toEmbodiment 4 of the present invention. It is to be noted that theoscillator is adapted to output a second harmonic of an oscillationsignal.

In FIG. 13, a collector of an emitter-grounded HBT 21 (first transistor)is connected to an emitter of a base-grounded HBT 22 (secondtransistor). In other words, the HBT 21 and the HBT 22 arecascode-connected. A collector of the HBT 22 is connected to an outputterminal.

A base of the HBT 22 is connected to a phase adjusting line 23 (signalimproving circuit) for adjusting the phase of an oscillation signal to adesired phase. An open stub 24 for short-circuiting high-frequencysignals in a predetermined frequency including the vicinity of thefrequency is connected to a side of the phase adjusting line 23 which isopposite to the HBT 22. Here, the length of the open stub 24 is set tobe ¼ the wavelength (λ/4) of an oscillation signal the oscillationfrequency of which is the predetermined frequency. It is to be notedthat, if necessary, an open stub the length of which is λ/4 of aharmonic of an oscillation signal may be additionally connected to thebase of the HBT 22.

Divider resistances 25 and 26 for setting a base voltage of the HBT 22are connected between an emitter of the HBT 21 and the base of the HBT22 and between the base and the collector of the HBT 22.

Further, a base of the HBT 21 is connected to a base bias circuit 27, afirst line 28, and a first stub 29. A collector bias circuit 30, asecond line 31, and a second stub 32 are connected between the collectorof the HBT 22 and the output terminal. It is to be noted that the firstline 28 and the first stub 29 form a resonant circuit.

The emitter of the HBT 21 is grounded through a third line 33.

The base bias circuit 27 and the collector bias circuit 30 havefunctions similar to those of the gate bias circuit 8 and the drain biascircuit 11, respectively, illustrated in FIG. 9.

The first stub 29 and the second stub 32 are a short stub and an openstub, respectively each length of which is λ/4 of an oscillation signal.By total reflection of an oscillation signal by these stubs, oscillationis made to grow.

With regard to a fundamental wave which is an oscillation signal,because it is reflected by the second stub 32, it is not output from theoutput terminal. With regard to a second harmonic of an oscillationsignal, because it is open to the second stub 32, it is not affected bythe second stub 32. Therefore, a second harmonic of an oscillationsignal is output from the output terminal.

The first line 28, the second line 31, and the third line 33 are linesprovided for adjusting the reflection gain and a reflection phase, andthe lengths of the lines are set so as to satisfy oscillationconditions.

Here, let an impedance when the side of the transistor is seen from asurface A-A of FIG. 13 (on the right of FIG. 13) be Z_(tr) and animpedance when the side of the resonant circuit is seen from the surfaceA-A (on the left of FIG. 13) be Z_(res). Generally, the oscillationconditions are satisfied and oscillation occurs when the frequencysatisfies both of the following Equations (1) and (2).

Re(Z _(tr))+Re(Z _(res))<0 and Re(Z _(tr))<0  (1)

Im(Z _(tr))+Im(Z _(res))=0  (2)

FIG. 14 is an explanatory graph of frequency characteristics ofRe(Z_(tr))+Re(Z_(res)) in the oscillator illustrated in FIG. 13. FIG. 15is an explanatory graph of frequency characteristics ofIm(Z_(tr))+Im(Z_(res)) in the oscillator illustrated in FIG. 13.

With reference to FIGS. 14 and 15, Equation (1) is satisfied at afrequency band of about 20 to about 50 GHz, and Equation (2) issatisfied at about 38 GHz. Therefore, in the oscillator according toEmbodiment 4, an oscillation signal at about 38 GHz is generated.

Next, the divider resistances 25 and 26 are described in detail.

FIG. 16 is an explanatory graph showing the relationship between aresistance ratio k of the divider resistance 25 and the dividerresistance 26 (k: resistance value of divider resistance 26/resistancevalue of divider resistance 25) and output power in the oscillatorillustrated in FIG. 13 with regard to a fundamental wave and a secondharmonic of an oscillation signal.

In FIG. 16, with regard to the fundamental wave, the output power is atthe maximum when k is 0.7, while, with regard to the second harmonic,the output power becomes higher as k becomes smaller. Therefore, inEmbodiment 4, k is set to be 0.1 to take out the second harmonic. It isto be noted that, when the fundamental wave is taken out, k may be setto be 0.7.

Next, the phase adjusting line 23 and the open stub 24 are described indetail.

As described above, the length of the open stub 24 is λ/4 of anoscillation signal, and an oscillation signal is short-circuited by thebase of the HBT 22.

The phase adjusting line 23 adjusts the phase of a second harmonic to adesired phase in order to optimize the output power of a second harmonicof an oscillation signal.

FIG. 17 is an explanatory graph showing the relationship between anelectrical length φ of the phase adjusting line 23 at the oscillationfrequency (38 GHz) and maximum output power of the second harmonic (76GHz) of an oscillation signal in the oscillator illustrated in FIG. 13.

The maximum output power of a second harmonic is output power of thesecond harmonic at an optimum load impedance with regard to therespective electrical lengths. As an example, FIG. 18 is a contour mapillustrating an output power of the second harmonic in the oscillatorillustrated in FIG. 13 when the electrical length φ is 18°. It can beseen from FIG. 18 that maximum output power of 21 dBm can be obtained atabout the center of the contour map. FIG. 17 plots maximum output powerof the contour map with respect to the respective electrical lengths.

From FIG. 17, even when the phase adjusting line 23 is not connected(more specifically, φ=0°), high output of about 15 dBm can be obtainedat the maximum. However, by shifting the phase of the second harmonic ofthe oscillation signal from the base of the HBT 22 by about 15-24°, theoutput power of the second harmonic can further be made higher.Therefore, in Embodiment 4, the electrical length φ is set to be 18°.

FIG. 19 is an explanatory graph of distribution of harmonics(oscillation spectrum) with regard to output power in the oscillatorillustrated in FIG. 13. In FIG. 19, the horizontal axis represents theorder of the harmonic. For example, 1 means a fundamental wave and 2means a second harmonic.

From FIG. 19, it can be seen that, in the oscillator which oscillates atabout 38 GHz, the output power of a second harmonic of an oscillationsignal is 21 dBm.

In this way, by using the phase adjusting line 23 and the open stub 24,a negative resistance |Re(Z_(tr))| can be made satisfactory high, and,because the output power of the second harmonic of the oscillationsignal can be optimized, the output power can be increased at themaximum.

In the oscillator according to Embodiment 4 of the present invention,the first transistor and the second transistor are cascode-connected,and the base of the second transistor is connected to the phaseadjusting line for adjusting the phase of an input signal to a desiredphase. The open stub for short-circuiting high-frequency signals in thepredetermined frequency including the vicinity of the frequency isconnected to the side of the phase adjusting line which is opposite tothe second transistor.

Therefore, a high-output oscillator at the millimeter-wave band can beobtained.

Further, because the negative resistance |Re(Z_(tr))| is determined notby the capacitance of a capacitor which varies greatly incharacteristics but by the length of the stub, manufacturing variationsof the oscillator can be suppressed.

It is to be noted that, although, in Embodiment 4, only the open stub 24the length of which is λ/4 of an oscillation signal is connected to theside of the phase adjusting line 23 which is opposite to the HBT 22, thepresent invention is not limited thereto, and an open stub for a secondharmonic (open stub for nth harmonic) the length of which is λ/4 of asecond harmonic of an oscillation signal may be further connected.Further, together with the phase adjusting line 23, a phase adjustingline for adjusting the phase of an oscillation signal to a desired phasemay be connected.

In this case, the output power of the second harmonic of the oscillationsignal can be further improved.

When the frequency to be used ranges over a plurality of bands,similarly to the case of Embodiment 1, an appropriate open stub may beselected among a plurality of open stubs using the selector switch 41(see FIG. 4), or, a variable length stub the length of which can beadjusted by the MEMS switch 42 may be used to change the frequency bandto be used (see FIG. 5).

Embodiment 5

Although, in Embodiment 4, using the open stub 24, the oscillationsignal is short-circuited by the base of the HBT 22, the presentinvention is not limited thereto.

In the following, a structure in which an oscillation signal isshort-circuited using a short stub is described. It is to be noted that,instead of the open stub or the short stub, a series resonant circuitadapted to resonate at the oscillation frequency of an oscillationsignal may be used.

FIG. 20 is a circuit diagram illustrating an oscillator according toEmbodiment 5 of the present invention. It is to be noted that theoscillator is adapted to output a second harmonic of an oscillationsignal. Description of structures and functions similar to thosedescribed in Embodiment 4 is omitted.

In FIG. 20, a base of an HBT 22 is connected to one end of a short stub34 (filter circuit) instead of the open stub 24 illustrated in FIG. 13.The short stub 34 short-circuits high-frequency signals in apredetermined frequency including the vicinity of the frequency. Here,the length of the short stub 34 is set to be ½ the wavelength (λ/2) ofan oscillation signal the oscillation frequency of which is thepredetermined frequency. It is to be noted that the length of the shortstub 34 may be set to be λ/2 of a harmonic of an oscillation signal.

The other end of the short stub 34 is grounded through avariable-capacitance capacitor 35 for filtering out a direct currentsignal.

It is to be noted that, because the oscillation frequency of theoscillator varies depending on the capacitance of the capacitorconnected to the other end of the short stub 34, by using thevariable-capacitance capacitor 35, the oscillation frequency can be madevariable.

Further, as the variable-capacitance element, a diode, for example, maybe used instead of the capacitor. The diode can be regarded as a circuitin which a variable-capacitance capacitor and a variable resistance areconnected in parallel as an equivalent circuit diagram illustrated inFIG. 21.

When a reverse voltage is applied to the diode, the resistance becomesseveral kiloohms or more, and thus, a high-frequency signal is greatlyaffected by a variable capacitance. Therefore, when a diode is used asthe variable-capacitance element, a voltage controlled oscillator whichcan control the oscillation frequency by the voltage applied to thediode can be formed.

Here, FIG. 22 is a circuit diagram illustrating a voltage controlledoscillator as the oscillator illustrated in FIG. 20 formed using a diodeas the variable-capacitance element.

In FIG. 22, a base of an HBT 22 is connected to a capacitor 36 and adiode 37 instead of the variable-capacitance capacitor 35 illustrated inFIG. 20. It is to be noted that the capacitor 36 is connected for thepurpose of filtering out a direct current signal.

An oscillation frequency control bias terminal to which a signal forcontrolling the oscillation frequency is input is connected between thecapacitor 36 and the diode 37.

In a conventional voltage controlled oscillator, a capacitor (varactor)for controlling the oscillation frequency as illustrated in FIG. 23 isconnected to a point Q of FIG. 22. However, in this case, there is aproblem that an impedance Re(Z_(res)) when the side of the resonantcircuit is seen from a surface A-A of FIG. 22 (on the left of FIG. 13)becomes higher and it is difficult that Equation (1) of theabove-mentioned oscillation conditions is satisfied.

In Embodiment 5, by connecting the varactor to the base of the HBT 22, avoltage controlled oscillator having a simpler circuit structure can beobtained without increasing the impedance Re(Z_(res)).

In the oscillator according to Embodiment 5 of the present invention,the first transistor and the second transistor are cascode-connected,and the base of the second transistor is connected to the phaseadjusting line for adjusting the phase of an input signal to a desiredphase. The open stub for short-circuiting high-frequency signals in thepredetermined frequency including the vicinity of the frequency isconnected to the side of the phase adjusting line which is opposite tothe second transistor.

Therefore, a high-output oscillator at the millimeter-wave band can beobtained.

Further, by grounding the other end of the short stub through the diode,a voltage controlled oscillator having a simple circuit structure can beobtained.

It is to be noted that, although, in Embodiment 5, only the short stub34 the length of which is λ/2 of an oscillation signal is connected tothe side of the phase adjusting line 23 which is opposite to the HBT 22,the present invention is not limited thereto, and a short stub for asecond harmonic (open stub for nth harmonic) the length of which is λ/2of a second harmonic of an oscillation signal may be further connected.Further, together with the phase adjusting line 23, a phase adjustingline for adjusting the phase of an oscillation signal to a desired phasemay be connected.

Instead of the short stub for a second harmonic, an open stub for asecond harmonic the length of which is λ/4 of a second harmonic of anoscillation signal may be connected together with a corresponding phaseadjusting line.

In those cases, the output power of the second harmonic of theoscillation signal can be further improved.

Further, when the impedance shifts from its optimum value by connectingthe varactor and the capacitor 36, the length of the short stub 34 maybe appropriately changed from λ/2 such that the output becomes optimum.

1. A cascode circuit including two cascode-connected transistors, comprising: a first transistor including, as a first terminal, one of a source and an emitter, the first terminal being grounded, and, as a second terminal, one of a drain and a collector; a second transistor including, as a first terminal, one of a source and an emitter, the first terminal being connected to the second terminal of the first transistor, as a second terminal, one of a gate and a base, and, as a third terminal, one of a drain and a collector; a signal improving circuit connected to the second terminal of the second transistor, for improving and outputting an input signal input to the signal improving circuit; and a filter circuit connected to a side of the signal improving circuit which is opposite a side of the signal improving circuit that is connected to the second transistor, for short-circuiting high-frequency signals at a predetermined frequency, including at frequencies near the predetermined frequency.
 2. The cascode circuit according to claim 1, wherein the filter circuit comprises a series resonant circuit including an inductor and a capacitor connected in series.
 3. The cascode circuit according to claim 1, wherein the filter circuit comprises an open stub.
 4. The cascode circuit according to claim 3, wherein the open stub has a length substantially ¼ of the wavelength of a signal at the predetermined frequency.
 5. The cascode circuit according to claim 3, wherein the open stub comprises a variable length stub having a length which can be adjusted by a micro electro mechanical system switch.
 6. The cascode circuit according to claim 3, further comprising: a plurality of the open stubs; and a selector switch for selecting one of the open stubs to be used from among the plurality of open stubs.
 7. The cascode circuit according to claim 1, wherein the filter circuit comprises a short stub.
 8. The cascode circuit according to claim 6, wherein the short stub has a length substantially ½ of the wavelength of a signal at the predetermined frequency.
 9. The cascode circuit according to claim 7, further comprising: a plurality of the short stubs; and a selector switch for selecting one of the short stubs to be used from among the plurality of short stubs.
 10. The cascode circuit according to claim 1, wherein the second transistor comprises a multistage transistor.
 11. The cascode circuit according to claim 1, wherein the first transistor and the second transistor comprise a dual-gate high electron mobility transistor.
 12. The cascode circuit according to claim 1, further comprising: a first diode including an anode that is grounded and a cathode connected to the second terminal of the second transistor; and a second diode including an anode connected to the second terminal of the second transistor and a cathode connected to the third terminal of the second transistor.
 13. The cascode circuit according to claim 1, wherein the signal improving circuit comprises a resistor for restricting reflection gain.
 14. The cascode circuit according to claim 1, wherein the signal improving circuit comprises a phase adjusting line for adjusting phase of the input signal.
 15. The cascode circuit according to claim 14, further comprising an open stub for an nth harmonic which has a length of substantially ¼ of the wavelength of an nth (n is an integer and at least 2) harmonic of a signal at the predetermined frequency and is connected to a side of the phase adjusting line which is opposite a side of the phase adjusting circuit connected to the second transistor.
 16. The cascode circuit according to claim 14, further comprising a short stub for an nth harmonic which has a length of substantially ½ of the wavelength of an nth (n is an integer and at least 2) harmonic of a signal at the predetermined frequency and is connected to a side of the phase adjusting line which is opposite a side of the phase adjusting circuit connected to the second transistor. 